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Dual-polarized antenna elements are widely used in such applications as polarization diversity [1] and radar target discrimination [2][3]. Dual-polarized square waveguide antennas are useful as a reflector feed and an array element [4]. Narrow-band excitation of a square waveguide can be done using dual coaxial probes [5] or a gap-fed patch [6], while wide-band operation requires quadruple ridges [7] and an orthomode transducer (OMT) [8]. In [6] a dual-polarized feeding method has been proposed employing a microstrip patch for a square horn. The coaxial probe feeding is simplest of all but its port isolation is poor. The gap-fed patch offers a good port isolation over a very narrow bandwidth [6]. Use of quadruple ridges or an OMT leads to wide bandwidth as well as high port isolation. Their principal disadvantages are structural complexity and large size.

In this paper, a new wideband square waveguide antenna fed by a microstrip patch is presented. Wideband dual-polarized characteristics have been obtained by feeding a cavity-backed square microstrip patch with two orthogonal coplanar-waveguide transmission lines through proximity coupling. The patch then excites the orthogonal TE₀₁ and TE₁₀ modes in the square waveguide. In the following, detailed design of the proposed antenna is presented. The CST Studio Suite 2019 has been employed in the antenna design.

Fig. 1 shows the structure of the proposed antenna where constituents are disassembled to help the reader's understanding.

Fig. 1. 
Structure of the proposed antenna

The proposed antenna consists of a square waveguide (A), a printed circuit board (B, B') containing a square microstrip patch on the front side and two orthogonal coplanar-waveguide transmission lines on the back side, and a metal plate (C, C') containing the air-filled cavity. Fig. 2 shows plane views of faces of the printed circuit board (B, B') and the metal plate (C).

Fig. 2. 
Plane views of faces B, B' and C of Fig. 1

Table 1 shows the design requirements of the antenna. The required bandwidth is not obtainable with a direct-fed microstrip patch. The bandwidth is greatly increased by proximity feeding the microstrip patch and by backing the patch and by backing the patch with a cavity.

The first step in the design of the proposed antenna is the determination of the square waveguide cross section. A large cross section results in the excitation of higher-order modes while a small one leads to a narrow bandwidth. A good choice of the square waveguide's width and height for the desired frequency 35-37 GHz is 7.20 mm. With this choice, the cutoff of the TE₁₀ (TE₀₁) mode occurs at 20.83 GHz, while the TE₂₀ (TE₀₂) is cut off at 41.66 GHz. The TE₁₁ mode with its cutoff frequency of 29.46 GHz is strongly suppressed due to the even symmetry in the waveguide excitation.

The next step in the design is the choice of the substrate. The Rogers RD/duroid 5870 substrate with ϵ_r = 2.33, tanδ = 0.002, h = 0.45 mm, t = 0.035 mm has been selected. Next a printed circuit transmission line feeding the patch is selected. Among many forms of planar transmission lines, a coplanar waveguide with a via-hole fence has been selected for its advantages such as suppression of the surface-wave coupling (thus high port-to-port isolation) and easy mounting of circuit elements of the RF front-end module. Fig. 3 shows the cross section of the printed circuit board containing a microstrip patch on the top face and the CPW feed lines on the bottom face.

Fig. 3. 
Cross section of the printed circuit board

The final step in the antenna design is the determination of the size of the cavity backing the square patch. The function of the cavity is the widening of the bandwidth and the suppression of the backward radiation from the patch. The cavity width and height determine the cutoff frequencies of the TE₁₀ (TE₀₁) and TE₂₀ (TE₀₂) of the cavity. Good performance is obtained when the cavity width and height are same as those of the square waveguide inner wall. The depth of the cavity is kept as small as possible for the required antenna bandwidth. A larger depth results in higher-order mode excitation leading to poor performance in port-to-port isolation and polarization discrimination.

Fig. 4 shows the important dimensions of the proposed antenna and Table 2 shows their values. The waveguide inner and outer wall dimensions are same as W₃ and W₂ respectively while its height is 8.0 mm. The cavity depth is 0.20 mm, which is same as the depth of the recessed region in the face C of Fig. 1

Fig. 4. 
Dimensional parameters of the proposed antenna

The final design of the antenna is obtained by a combination of theory, parameter sweep and optimization. The most critical dimensions in the antenna design are the feed gap g and the waveguide wall width W₃.

Fig. 5 shows the input reflection coefficient versus the variations in the feed gap. The feed gap g is fine tuned to obtain a reflection coefficient of less than -10 dB at 35-37 GHz. The feed gap has minor effects on the port-to-port coupling. With all other dimensions fixed, the waveguide wall width W₃ has sensitive effects on the reflection and port-to-port coupling as shown in Fig. 6.

Fig. 5. 
Reflection coefficient versus feed gap

Fig. 6. 
Effects of the waveguide width on (a) the reflection and (b) the port-to-port coupling

Fig. 7 shows the electric field distribution on the CPW feed line. The characteristic impedance of the line is 50 Ω. Fig. 8 shows the electric field distribution around the patch when one of the feed line is excited while the other is terminated with a matched load. One can observe a good polarization uniformity. Fig. 9 shows the corresponding electric field distribution at a cross section of the square waveguide. Again the polarization purity is high containing only the TE₁₀ mode field.

Fig. 7. 
Electric field on a cross section of the feed line

Fig. 8. 
Electric field around the microstrip patch

Fig. 9. 
Electric field on the square waveguide cross section

Fig. 10 shows the port reflection coefficient and the port-to-port coupling coefficient (or the negative of the port isolation). At 35-37 GHz, the reflection and port-to-port coupling are less than -10 dB and -24 dB respectively. Fig. 11 shows the three-dimensional gain patterns of the designed antenna at 35 GHz. One can observe a reasonably good angular symmetry in the gain pattern. Figs. 12-13 show the co-polarization gain in the E- and H-planes and the cross-polarization gain in the diagonal planes at 35 and 37 GHz. For the co-polarized radiation, the gain ranges from 9.5 to 9.9 dBi while the cross polarization at boresight ranges from -25 to -18 dBi resulting in cross polarization discrimination greater then 20 dB. The front-to-back ratio is 28 dB at 35 GHz and 35 dB at 37 GHz. The final design of the proposed antenna meets all the design requirements in Table 1.

Fig. 10. 
Reflection and coupling coefficients of the designed antenna

Fig. 11. 
3D gain pattern of the designed antenna

Fig. 12 
Gain patterns of the designed antenna at 35 GHz

Fig. 13. 
Gain patterns of the designed antenna at 35 GHz

A dual-polarized square waveguide antenna has been designed that has good characteristics at 35-37 GHz. A proximity-coupled dual-CPW fed square patch has been designed to excite the orthogonal TE₁₀ and TE₀₁ modes in the square waveguide. The bandwidth limitation of the patch antenna has been solved by feeding the patch with proximity coupling from the back side and by backing the patch with an air-filled cavity. At desired 35-37 GHz frequency range, the designed antenna has reflection coefficient of less than -10 dB, port-to-port isolation of greater than 24 dB, and boresight cross-polarization discrimination of greater than 28 dB. The antenna design presented in this paper may be employed as a dual-polarized reflector antenna feed and a radiating element in a dual-polarized waveguide array antenna.